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MIC2169B
500kHz PWM Synchronous Buck Control IC
General Description
The MIC2169B is a high-efficiency, simple to use 500kHz PWM synchronous buck control IC housed in a small MSOP-10 ePad package. It allows compact DC/DC solutions with a minimal external component count and cost. The device features high output driver capability to drive loads up to 30A. The MIC2169B operates from a 3V to 14.5V input, without the need of any additional bias voltage. The output voltage can be precisely regulated down to 0.8V. The adaptive all N-Channel MOSFET drive scheme allows efficiencies over 95% across a wide load range within the smallest possible printed circuit board space area. The MIC2169B senses current across the high-side NChannel MOSFET, eliminating the need for an expensive and lossy current-sense resistor. Current limit accuracy is maintained by a positive temperature coefficient that tracks the increasing RDS(ON) of the external MOSFET. Further cost and space are saved by the internal in-rush-current limiting digital soft-start. The MIC2169B is identical to the MIC2169A except it supports pre-bias loads. Internal prebias circuit prevents output voltage drooping and excessive reverse inductor current when powering up with a pre-bias voltage at the output. The MIC2169B is available in a thermally capable 10-pin ePad MSOP package, with a wide junction operating range of -40C to +125C. All support documentation can be found on Micrel's web site at www.micrel.com.
Features
* * * * * * * * * * * * * * * * * * * * * * * 3V to 14.5V input voltage range Adjustable output voltage down to 0.8V 500kHz PWM operation Up to 95% efficiency Output Pre-biased Protection Build-in 2.2 drivers to drive two N-channel MOSFETs Adaptive gate drive increases efficiency Simple voltage-mode PWM control with externally compensated Short minimum ON time of 30ns allowing very low duty cycle Fast transient response Adjustable current limit senses high-side N-Channel MOSFET current Hiccup mode short-circuit protection No external current-sense resistor Internal soft-start current source Dual function COMP and EN pin allows low-power shutdown Available in a small size 10-pin MSOP ePad package Point-of-load DC/DC conversion High Current Power Supplies Telecom/Datacom and Networking Power Supplies Servers and Workstations Graphic cards and other PC Peripherals Set-top boxes LCD power supplies
Applications
___________________________________________________________________________________________________________
Micrel Inc. * 2180 Fortune Drive * San Jose, CA 95131 * USA * tel +1 (408) 944-0800 * fax + 1 (408) 474-1000 * http://www.micrel.com
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MIC2169B
Typical Application
VIN = 5V to 12V 100F 10F SD103BWS 10F 0.1F
VDD
BST CS IRF7821 1.0H IRF7821 1000pF 330F x 2 3.3V
VIN
HSD MIC2169B VSW LSD
EFFICIENCY (%)
100 95 90 85 80 75 70 65 60 55 50
MIC2169B Efficienc
COMP/EN 150pF 100nF 1F GND EP FB
VIN = 5V VOUT = 3.3V 0 2 4 6 8 10 12 14 16 ILOAD (A)
MIC2169B Adjustable Output 500kHz Converter
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MIC2169B
Ordering Information
Part Number MIC2169BYMME Frequency 500kHz Junction Temp. Range(1) -40 to +125C Package 10-Pin ePad MSOP Lead Finish Pb-Free
Pin Configuration
VIN 1 VDD 2 CS 3 COMP/EN 4 FB 5 EP
10 BST 9 8 7 6 HSD VSW LSD GND
10-Pin ePad MSOP (MME)
Pin Description
Pin Number Pin Name Pin Function
1 2
VIN VDD
Supply Voltage (Input): +3V to +14.5V. 5V Internal Linear Regulator (Output): VDD is the external MOSFET gate drive supply voltage and an internal supply bus for the IC. When VIN is <5V, short VDD to the input supply through a 10 resistor. Current Sense (Input): Current-limit comparator noninverting input. The current limit is sensed across the MOSFET during the ON time. The current can be set by the resistor in series with the CS pin. Compensation / Enable (Input): Dual function pin. Pin for external compensation. If this pin is pulled below 0.25V, with the reference fully up the device shuts down (50A typical current draw). Feedback (Input): Input to error amplifier. Regulates error amplifier to 0.8V. Ground (Return). Low-Side Drive (Output): High-current driver output for external synchronous MOSFET. Switch (Return): High-side MOSFET driver return. High-Side Drive (Output): High-current output-driver for the high-side MOSFET. When VIN is between 3.0V to 5V, 2.5V threshold MOSFETs should be used. At VIN > 5V, 4.5V threshold MOSFETs should be used. Boost (Input): Provides the drive voltage for the high-side MOSFET driver. The gate-drive voltage is higher than the source voltage by VDD minus a diode drop. Connect to Ground
3
CS
4
COMP/EN
5 6 7 8 9
FB GND LSD VSW HSD
10 ePad
BST EP
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Micrel, Inc.
MIC2169B
Absolute Maximum Ratings(1)
Supply Voltage (VIN) ...................................... -0.3V to 15.5V Booststrapped Voltage (VBST) .................... -0.3V to VIN +6V VSW .............................................................. -0.3V to 15.5V CS ............................................................................15.25V FB ..................................................................... -0.3V to 6V Junction Temperature (TJ) .................. -40C TJ +125C Storage Temperature (TS)..........................-65C to +150C Peak Reflow Temperature (10 to 20 sec) ................ +260C ESD (HBM) (3) ................................................................. 2kV ESD (MM).....................................................................200V
Operating Ratings(2)
Supply Voltage (VIN)...................................... +3V to +14.5V Ambient Temperature (TA) .........................-40C to +125C Junction Thermal Resistance ePad MSOP (JA)............................................76.7C/W Output Voltage Range............................. 0.8V to VIN x DMAX
Electrical Characteristics(4)
TJ = 25C, VIN = 5V; bold values indicate -40C TJ +125C; unless otherwise specified. Parameter Feedback Voltage Reference Feedback Voltage Reference Feedback Bias Current Output Voltage Line Regulation Output Voltage Load Regulation Output Voltage Total Regulation Oscillator Section Oscillator Frequency Maximum Duty Cycle Minimum On-Time
(5)
Condition (1%) (2% over temp)
Min 0.792 0.784
Typ 0.8 0.8 150 0.03 0.5
Max 0.808 0.816 350
Units V V nA %/V %
3V VIN 14.5V; 1A IOUT 10A; (VOUT = 2.5V)(4)
0.6
1.5
%
450 92
500 30
550 60 3 150 0.35
kHz % ns mA A V s
Input and VDD Supply PWM Mode Supply Current Shutdown Quiescent Current VCOMP Shutdown Threshold VCOMP Shutdown Blanking Period Digital Supply Voltage (VDD) Error Amplifier DC Gain(5) Transconductance 70 1.1 dB m-1 CCOMP = 100nF VIN 6V 4.7 VCS = VIN -0.25V; VFB = 0.7V (output switching but excluding external MOSFET gate current.) VCOMP/EN = 0V 0.1 1.5 50 0.25 675 5 5.3
V
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MIC2169B
Parameter Soft-Start Soft-Start Current Current Sense CS Over Current Trip Point Temperature Coefficient Gate Drivers Rise/Fall Time Output Driver Impedance
Condition After time out of internal timer. VCOMP = 0.8V VCS = VIN -0.25V
Min 4 160
Typ 8.5 200 1800
Max 13 240
Units A A ppm/C ns
Into 3000pF at VIN > 5V Source, VIN = 4.5V Sink, VIN = 4.5V Source, VIN = 3V Sink, VIN = 3V
15
2.2 1.3 2.7 1.7
3 3 4 4
ns
Driver Non-Overlap Time
Notes:
(5)
50
1. Absolute maximum ratings indicate limits beyond which damage to the component may occur. Electrical specifications do not apply when operating the device outside of its operating ratings. The maximum allowable power dissipation is a function of the maximum junction temperature, TJ(max), the junction-to-ambient thermal resistance, JA, and the ambient temperature, TA. The maximum allowable power dissipation will result in excessive die temperature. 2. The device is not guaranteed to function outside its operating rating. 3. Devices are ESD sensitive, handling precautions required. 4. Specification for packaged product only. 5. Guaranteed by design.
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MIC2169B
Typical Characteristics
2.9 2.7 2.5 2.3 2.1 1.9 1.7 1.5 1.3 1.1 0.9 0.7 0.5 -40 -20 0 20 40 60 80 100120140 TEMPERATURE (C)
PWM Mode Supply Current vs. Temperature
2.0 QUIESCENT CURRENT (mA)
PWM Mode Supply Current vs. Suppl Voltage
0.8010 0.8005
VFB Line Regulation
IDD (mA)
VFB (V)
1.5
0.8000 0.7995 0.7990 0.7985
1.0
0.5
0
5 10 SUPPLY VOLTAGE (V)
15
0.7980
0
5 VIN (V)
10
15
0.804 0.802 VFB (V) 0.800 0.798 0.796 0.794 0.792 -60 -30 0 30 60 90 120 150 TEMPERATURE (C)
VDD (V)
VDD REGULATOR VOLTAGE (V)
0.806
VFB vs. Temperature
6 5 4 3 2 1 0 0
VDD Line Regulation
5.02 5.00 4.98 4.96 4.94 4.92 4.90 0
VDD Load Regulation
5 VIN (V)
10
15
5 10 15 20 25 LOAD CURRENT (mA)
30
5.0 VDD LINE REGULATION (%) 4.5 4.0 3.5 3.0 2.5 2.0 1.5 1.0 0.5
VDD Line Regulation vs. Temperature
FREQUENCY VARIATION (%)
FREQUENCY (kHz)
550 540 530 520 510 500 490 480
Oscillator Frequency vs. Temperature
1.5 1.0 0.5 0 -0.5 -1.0 -1.5 0
Oscillator Frequency vs. Suppl Voltage
0.0 -60 -30 0 30 60 90 120 150 TEMPERATURE (C)
470 460 450 -60 -30 0 30 60 90 120 150 TEMPERATURE (C)
5 VIN (V)
10
15
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Overcurrent Trip Point vs. Temperature
MIC2169B
260 240 220 ICS ( A) 200 180 160 140 120
100 -60 -30 0 30 60 90 120 150 TEMPERATURE (C)
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MIC2169B
Functional Diagram
MIC2169B Block Diagram
Functional Description
The MIC2169B is a voltage mode, synchronous stepdown switching regulator controller designed for high power. Current limit is implemented without the use of an external sense resistor. It includes an internal soft-start function which reduces the power supply input surge current at start-up by controlling the output voltage rise time, a PWM generator, a reference voltage, two MOSFET drivers, and short-circuit current limiting circuitry to form a complete 500kHz switching regulator. MIC2169B is identical to the MIC2169A except it supports pre-bias loads. Theory of Operation The MIC2169B is a voltage mode step-down regulator. The figure above illustrates the block diagram for the voltage control loop. The output voltage variation due to June 2009 8
load or line changes will be sensed by the inverting input of the transconductance error amplifier via the feedback resistors R3, and R2 and compared to a reference voltage at the non-inverting input. This will cause a small change in the DC voltage level at the output of the error amplifier which is the input to the PWM comparator. The other input to the comparator is a 0.95V to 1.45V triangular waveform. The comparator generates a rectangular waveform whose width tON is equal to the time from the start of the clock cycle t0 until t1, the time the triangle crosses the output waveform of the error amplifier. To illustrate the control loop, let us assume the output voltage drops due to sudden load turn-on, this would cause the inverting input of the error amplifier, which is divided down version of VOUT, to be slightly less than the reference voltage, causing the output voltage of the error amplifier to go high. This will cause the PWM comparator to increase tON time of the top side
M9999- 060309-A (408) 944-0800
Micrel, Inc. MOSFET, causing the output voltage to go up and bringing VOUT back in regulation. Soft-Start The COMP/EN pin on the MIC2169B is used for the following three functions: 1. Disables the part by grounding this pin 2. External compensation to stabilize the voltage control loop 3. Soft-start For better understanding of the soft-start feature, assume VIN = 12V, and the MIC2169B is allowed to power-up by un-grounding the COMP/EN pin. The COMP pin has an internal 8.5A current source that charges the external compensation capacitor. As soon as this voltage rises to 250mV (t = Cap_COMP x 0.25V/8.5A) and VIN crosses the 2.6V UVLO threshold, the MIC2169B allows the internal VDD linear regulator to power up, and the chip's internal oscillator starts switching. At this point in time, the COMP pin current source increases to 40A and an internal 11-bit counter starts counting which takes approximately 2ms to complete. During counting, the COMP voltage is clamped at 0.65V. After this counting cycle the COMP current source is reduced to 8.5A and the COMP pin voltage rises from 0.65V to 0.95V, the bottom edge of the saw-tooth oscillator. This is the beginning of 0% duty cycle and it increases slowly causing the output voltage to rise slowly. The MIC2169B has one hysteretic comparator whose output is asserted high when VOUT is within -3% of steady state. When the output voltage reaches 97% of programmed output voltage then the gm error amplifier is enabled along with the hysteretic comparator output is asserted high. This point onwards, the voltage control loop (gm error amplifier) is fully in control and will regulate the output voltage. Soft-start time can be calculated approximately by adding the following four time frames: t1 = Cap_COMP x 0.25V/8.5A t2 = 12 bit counter, approx 2ms t3 = Cap_COMP x 0.3V/8.5A
V t 4 = OUT V IN Cap _ COMP x 0. 5 x 8.5A
MIC2169B not very accurate. This scheme is adequate to protect the power supply and external components during a fault condition by cutting back the time the top MOSFET is on if the feedback voltage is greater than 0.67V. In case of a hard short when feedback voltage is less than 0.67V, the MIC2169B discharges the COMP capacitor to 0.65V, resets the digital counter and automatically shuts off the top gate drive, the gm error amplifier is completely disabled, the -3% hysteretic comparators is asserted low, and the soft-start cycles restart from t2 to t4. This mode of operation is called the "hiccup mode" and its purpose is to protect the down stream load in case of a hard short. The circuit in Figure 1 illustrates the MIC2169B current limiting circuit.
VIN C2 CIN 0.1F L1 Inductor RCS VSW CS LSD Q2 MOSFET N 1000pF C1 COUT VOUT
HSD
Q1 MOSFET N
200 A
Figure 1. The MIC2169B Current Limiting Circuit
The current limiting resistor RCS is calculated by the following equation:
RCS = RDS(ON)Q1 x IL 200A Inductor Ripple Current 2
where:
IL = ILOAD +
Inductor Ripple Current = VOUT x
(VIN - VOUT )
VIN x FS x L
Soft-Start Time(Cap_COMP=100nF) = t1 + t2 + t3 + t4 = 2.9ms + 2ms + 3.5ms + 1.6ms = 10ms Current Limit The MIC2169B uses the RDS(ON) of the top power MOSFET to measure output current. Since it uses the drain to source resistance of the power MOSFET, it is
FS = 500kHz 200A is the internal sink current to program the MIC2169B current limit. The MOSFET RDS(ON) varies 30% to 40% with temperature; therefore, it is recommended to add a 50% margin to the load current (ILOAD) in the above equation to avoid false current limiting due to increased MOSFET junction temperature rise. It is also recommended to connect RCS resistor directly to the drain of the top MOSFET Q1, and the RSW resistor to the source of Q1 to accurately sense the MOSFETs RDS(ON). To make the MIC2169B insensitive to board layout and noise generated by the switch node, a 1.4 resistor and a 9
M9999- 060309-A (408) 944-0800
June 2009
Micrel, Inc. 1000pF capacitor is recommended between the switch node and GND. Internal VDD Supply The MIC2169B controller internally generates VDD for self biasing and to provide power to the gate drives. This VDD supply is generated through a low-dropout regulator and generates 5V from VIN supply greater than 5V. For supply voltage less than 5V, the VDD linear regulator is approximately 200mV in dropout. Therefore, it is recommended to short the VDD supply to the input supply through a 10 resistor for input supplies between 3.0V to 5V. MOSFET Gate Drive The MIC2169B high-side drive circuit is designed to switch an N-Channel MOSFET. The Functional Block Diagram on page 8 shows a bootstrap circuit, consisting of D1 and CBST, supplies energy to the high-side drive circuit. Capacitor CBST is charged while the low-side MOSFET is on and the voltage on the VSW pin is
MIC2169B approximately 0V. When the high-side MOSFET driver is turned on, energy from CBST is used to turn the MOSFET on. As the MOSFET turns on, the voltage on the VSW pin increases to approximately VIN. Diode D1 is reversed biased and CBST floats high while continuing to keep the high-side MOSFET on. When the low-side switch is turned back on, CBST is recharged through D1. The drive voltage is derived from the internal 5V VDD bias supply. The nominal low-side gate drive voltage is 5V and the nominal high-side gate drive voltage is approximately 4.5V due the voltage drop across D1. An approximate 50ns delay between the high-side and low-side driver transitions is used to prevent current from simultaneously flowing unimpeded through both MOSFETs (shoot-through). Adaptive gate drive is implemented on the high-side (off) to low-side (on) driver transition to reduce losses in the flywheel diode and to prevent shoot-through. This is operated by detecting the VSW pin; once this pin is detected to reach 1.5V, the high-side MOSFET can be assumed to be off and the low side driver is enabled.
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MIC2169B A convenient figure of merit for switching MOSFETs is the on resistance times the total gate charge RDS(ON)xQG. Lower numbers translate into higher efficiency. Low gate-charge logic-level MOSFETs are a good choice for use with the MIC2169B. Parameters that are important to MOSFET switch selection are: * Voltage rating * On-resistance * Total gate charge The voltage ratings for the top and bottom MOSFET are essentially equal to the input voltage. A safety factor of 20% should be added to the VDS(max) of the MOSFETs to account for voltage spikes due to circuit parasitics. The power dissipated in the switching transistor is the sum of the conduction losses during the on-time (PCONDUCTION) and the switching losses that occur during the period of time when the MOSFETs turn on and off (PAC).
PSW = PCONDUCTION + PAC
Application Information
MOSFET Selection The MIC2169B controller works from input voltages of 3V to 14.5V and has an internal 5V regulator to provide power to turn the external N-Channel power MOSFETs for high- and low-side switches. For applications where VIN < 5V, the internal VDD regulator operates in dropout mode, and it is necessary that the power MOSFETs used are sub-logic level and are in full conduction mode for VGS of 2.5V. For applications when VIN > 5V; logiclevel MOSFETs, whose operation is specified at VGS = 4.5V must be used. For the lower (<5V) applications, the VDD supply can be connected directly to VIN to help increase the driver voltage to the MOSFET. It is important to note the on-resistance of a MOSFET increases with increasing temperature. A 75C rise in junction temperature will increase the channel resistance of the MOSFET by 50% to 75% of the resistance specified at 25C. This change in resistance must be accounted for when calculating MOSFET power dissipation and in calculating the value of current-sense (CS) resistor. Total gate charge is the charge required to turn the MOSFET on and off under specified operating conditions (VDS and VGS). The gate charge is supplied by the MIC2169B gate-drive circuit. At 500kHz switching frequency and above, the gate charge can be a significant source of power dissipation in the MIC2169B. At low output load, this power dissipation is noticeable as a reduction in efficiency. The average current required to drive the high-side MOSFET is:
IG[high -side](avg ) = Q G x f S
where:
PCONDUCTION = ISW (rms )2 x RSW PAC = PAC(off ) + PAC(on)
RSW = on-resistance of the MOSFET switch
V D = duty cycle = O V IN
where: IG[high-side](avg) = average high-side MOSFET gate current. QG = total gate charge for the high-side MOSFET taken from manufacturer's data sheet for VGS = 5V. The low-side MOSFET is turned on and off at VDS = 0 because the freewheeling diode is conducting during this time. The switching loss for the low-side MOSFET is usually negligible. Also, the gate-drive current for the low-side MOSFET is more accurately calculated using CISS at VDS = 0 instead of gate charge. For the low-side MOSFET:
IG[low - side](avg ) = CISS x VGS x fS
Making the assumption the turn-on and turn-off transition times are equal; the transition times can be approximated by:
tT = CISS x VGS + COSS x VIN IG
where: CISS and COSS are measured at VDS = 0 IG = gate-drive current (1.4A for the MIC2169B) The total high-side MOSFET switching loss is:
PAC = (VIN + VD ) x IPK x t T x fS
where: tT = switching transition time (typically 20ns to 50ns) VD = freewheeling diode drop, typically 0.5V fS it the switching frequency, nominally 500kHz The low-side MOSFET switching losses are negligible and can be ignored for these calculations.
Since the current from the gate drive comes from the input voltage, the power dissipated in the MIC2169B due to gate drive is:
PGATEDRIVE = VIN x IG[high -side](avg) + IG[low -side](avg)
(
)
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Micrel, Inc. Inductor Selection Values for inductance, peak, and RMS currents are required to select the output inductor. The input and output voltages and the inductance value determine the peak-to-peak inductor ripple current. Generally, higher inductance values are used with higher input voltages. Larger peak-to-peak ripple currents will increase the power dissipation in the inductor and MOSFETs. Larger output ripple currents will also require more output capacitance to smooth out the larger ripple current. Smaller peak-to-peak ripple currents require a larger inductance value and therefore a larger and more expensive inductor. A good compromise between size, loss and cost is to set the inductor ripple current to be equal to 20% of the maximum output current. The inductance value is calculated by the equation below.
L= VOUT x VIN(max) - VOUT
MIC2169B usually insignificant and can be ignored. At lower output currents, the core losses can be a significant contributor. Core loss information is usually available from the magnetics vendor. Copper loss in the inductor is calculated by the equation below:
PINDUCTORCu = IINDUCTOR(rms )2 x R WINDING
The resistance of the copper wire, RWINDING, increases with temperature. The value of the winding resistance used should be at the operating temperature.
R WINDING(hot ) = R WINDING( 20C) x (1 + 0.0042 x (THOT - T20C )
where: THOT = temperature of the wire under operating load T20C = ambient temperature RWINDING(20C) = room temperature winding resistance (usually specified by the manufacturer) Output Capacitor Selection The output capacitor values are usually determined by the capacitors ESR (equivalent series resistance). Voltage and RMS current capability are two other important factors selecting the output capacitor. Recommended capacitors are tantalum, low-ESR aluminum electrolytics, and POSCAPS. The output capacitor's ESR is usually the main cause of output ripple. The output capacitor ESR also affects the overall voltage feedback loop from stability point of view. See "Feedback Loop Compensation" section for more information. The maximum value of ESR is calculated:
RESR VOUT IPP
(
)
VIN(max) x f S x 0.2 x IOUT(max)
where:
fS = switching frequency, 500kHz 0.2 = ratio of AC ripple current to DC output current VIN(max) = maximum input voltage The peak-to-peak inductor current (AC ripple current) is:
IPP = VOUT x VIN(max) - VOUT VIN(max) x fS x L
(
)
The peak inductor current is equal to the average output current plus one half of the peak-to-peak inductor ripple current.
IPK = IOUT(max) + 0.5 x IPP
The RMS inductor current is used to calculate the I2 x R losses in the inductor.
I2 IINDUCTOR = (IOUT _ MAX ) 2 + PP 12
Maximizing efficiency requires the proper selection of core material and minimizing the winding resistance. The high frequency operation of the MIC2169B requires the use of ferrite materials for all but the most cost sensitive applications. Lower cost iron powder cores may be used but the increase in core loss will reduce the efficiency of the power supply. This is especially noticeable at low output power. The winding resistance decreases efficiency at the higher output current levels. The winding resistance must be minimized although this usually comes at the expense of a larger inductor. The power dissipated in the inductor is equal to the sum of the core and copper losses. At higher output loads, the core losses are June 2009 12
where: VOUT = peak-to-peak output voltage ripple IPP = peak-to-peak inductor ripple current The total output ripple is a combination of the ripple due to the output capacitors' ESR and the ripple due to the output capacitor. The total ripple is calculated below: VOUT I x (1 - D) + (IPP x RESR )2 = PP C OUT x fS
2
where: D = duty cycle COUT = output capacitance value fS = switching frequency The voltage rating of capacitor should be twice the voltage for a tantalum and 20% greater for aluminum electrolytic. The output capacitor RMS current is calculated below:
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I IC OUT ( rms ) = PP 12
MIC2169B
The power dissipated in the output capacitor is:
PDISS(C OUT ) = IC OUT ( rms )
(
)2 x RESR(C
the voltage feedback loop. If R1 is too small, in value, it will decrease the efficiency of the power supply, especially at light loads. Once R1 is selected, R2 can be calculated using:
R2 = VREF x R1 VO - VREF
OUT
)
Input Capacitor Selection The input capacitor should be selected for ripple current rating and voltage rating. Tantalum input capacitors may fail when subjected to high inrush currents, caused by turning the input supply on. A tantalum input capacitor's voltage rating should be at least 2 times the maximum input voltage to maximize reliability. Aluminum electrolytic, OS-CON, and multilayer polymer film capacitors can handle the higher inrush currents without voltage derating. The input voltage ripple will primarily depend on the input capacitor's ESR. The peak input current is equal to the peak inductor current, so:
VIN = IINDUCTOR(peak ) x RESR(CIN )
External Schottky Diode An external freewheeling diode is used to keep the inductor current flow continuous while both MOSFETs are turned off. This dead time prevents current from flowing unimpeded through both MOSFETs and is typically 50ns. The diode conducts twice during each switching cycle. Although the average current through this diode is small, the diode must be able to handle the peak current.
The input capacitor must be rated for the input current ripple. The RMS value of input capacitor current is determined at the maximum output current. Assuming the peak-to-peak inductor ripple current is low:
ICIN( rms ) IOUT(max) x D x (1 - D)
ID(avg) = IOUT x 2 x 50ns x fS The reverse voltage requirement of the diode is: VDIODE(rrm) = VIN The power dissipated by the Schottky diode is: PDIODE = ID(avg) x VF where: VF = forward voltage at the peak diode current The external Schottky diode, D1, is not necessary for circuit operation since the low-side MOSFET contains a parasitic body diode. The external diode will improve efficiency and decrease high frequency noise. If the MOSFET body diode is used, it must be rated to handle the peak and average current. The body diode has a relatively slow reverse recovery time and a relatively high forward voltage drop. The power lost in the diode is proportional to the forward voltage drop of the diode. As the high-side MOSFET starts to turn on, the body diode becomes a short circuit for the reverse recovery period, dissipating additional power. The diode recovery and the circuit inductance will cause ringing during the high-side MOSFET turn-on. An external Schottky diode conducts at a lower forward voltage preventing the body diode in the MOSFET from turning on. The lower forward voltage drop dissipates less power than the body diode. The lack of a reverse recovery mechanism in a Schottky diode causes less ringing and less power loss. Depending on the circuit components and operating conditions, an external Schottky diode will give a 1/2% to 1% improvement in efficiency.
Feedback Loop Compensation The MIC2169B controller comes with an internal transconductance error amplifier used for compensating the voltage feedback loop by placing a capacitor (C1) in series with a resistor (R1) and another capacitor C2 in parallel from the COMP pin to ground. See "Functional 13
M9999- 060309-A (408) 944-0800
The power dissipated in the input capacitor is:
PDISS(CIN ) = ICIN(rms )
(
)2 x RESR(C
IN
)
Voltage Setting Components The MIC2169B requires two resistors to set the output voltage as shown in Figure 2.
R1 Error Amp FB
5
R2 VREF 0.8V MIC2169B
Figure 2. Voltage-Divider Configuration
The output voltage is determined by the equation:
R1 VO = VREF x 1 + R2 where: VREF for the MIC2169B is typically 0.8V A typical value of R1 can be between 3k and 10k. If R1 is too large, it may allow noise to be introduced into
June 2009
Micrel, Inc. Block Diagram."
Power Stage The power stage of a voltage mode controller has an inductor, L1, with its winding resistance (DCR) connected to the output capacitor, COUT, with its electrical series resistance (ESR) as shown in Figure 3. The transfer function G(s), for such a system is:
L DCR VO ESR COUT
MIC2169B
Figure 5. Figure 3. The Output LC Filter in a Voltage Mode Buck Converter
Phase Curve for G(s)
(1 + ESR x s x C) G(s) = 2 DCR x s x C + s x L x C + 1 + ESR x s x C
It can be seen from the transfer function G(s) and the gain curve that the output inductor and capacitor create a two pole system with a break frequency at:
fLC = 1 2 x L x COUT
Plotting this transfer function with the following assumed values (L=1H, DCR=0.009, COUT=660F, ESR=0.025) gives lot of insight as to why one needs to compensate the loop by adding resistor and capacitors on the COMP pin. Figures 4 and 5 show the gain curve and phase curve for the above transfer function.
Therefore, fLC = 6.2kHz By looking at the phase curve, it can be seen that the output capacitor ESR (0.025) cancels one of the two poles (LCOUT) system by introducing a zero at:
fZERO = 1 2 x x ESR x COUT
Therefore, FZERO = 9.6kHz. From the point of view of compensating the voltage loop, it is recommended to use higher ESR output capacitors since they provide a 90 phase gain in the power path. For comparison purposes, Figure 6, shows the same phase curve with an ESR value of 0.002.
Figure 4.
The Gain Curve for G(s)
Figure 6.
The Phase Curve with ESR = 0.002
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Micrel, Inc. It can be seen from Figure 5 that at 50kHz, the phase is approximately -90 versus Figure 6 where the number is -150. This means that the transconductance error amplifier has to provide a phase boost of about 45 to achieve a closed loop phase margin of 45 at a crossover frequency of 50kHz for Figure 5, versus 105 for Figure 6. The simple RC and C2 compensation scheme allows a maximum error amplifier phase boost of about 90. Therefore, it is easier to stabilize the MIC2169B voltage control loop by using high ESR value output capacitors.
gm Error Amplifier It is undesirable to have high error amplifier gain at high frequencies because high frequency noise spikes would be picked up and transmitted at large amplitude to the output, thus, gain should be permitted to fall off at high frequencies. At low frequency, it is desired to have high open-loop gain to attenuate the power line ripple. Thus, the error amplifier gain should be allowed to increase rapidly at low frequencies. The transfer function with R1, C1, and C2 for the internal gm error amplifier can be approximated by the following equation:
1 + s x R1x C1 Error Amplifier (z) = gm x C1x C2 s x (C1 + C2) x 1 + s x R1x C1 + C2
MIC2169B
Figure 7.
Error Amplifier Gain Curve
The above equation can be simplified by assuming C2< 1 + s x R1 x C1 Error Amplifier(z) = gm x s x C1 x (1 + s x R1 x C2)
Figure 8.
Error Amplifier Phase Curve
From the above transfer function, one can see that R1 and C1 introduce a zero and R1 and C2 a pole at the following frequencies: FZERO= 1/2 x R1 x C1 FPOLE = 1/2 x C2 x R1 FPOLE@origin = 1/2 x C1 Figures 7 and 8 show the gain and phase curves for the above transfer function with R1 = 4.02k, C1 = 100nF, C2 = 150pF, and gm = 1.1m-1.
Total Open-Loop Response The open-loop response for the MIC2169B controller is easily obtained by adding the power path and the error amplifier gains together, since they already are in Log scale. It is desirable to have the gain curve intersect zero dB at tens of kilohertz, this is commonly called crossover frequency; the phase margin at crossover frequency should be at least 45. Phase margins of 30 or less cause the power supply to have substantial ringing when subjected to transients, and have little tolerance for component or environmental variations. Figures 9 and 10 show the open-loop gain and phase margin for the 5V input and 1.8V output application, and it can be seen from Figure 9 that the gain curve intersects the 0dB at approximately 50kHz, and from Figure 10 that at 50kHz, the phase shows approximately 74 of margin.
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MIC2169B
Figure 9.
Open-Loop Gain Margin
Figure 11. MIC2169A Startup without Pre-Bias Support
Figure 11 shows MIC2169A startup with a pre-bias of 1V on the output, in which the pre-existing output voltage discharges during soft start.
Figure 10.
Open-Loop Phase Margin
Pre-Biased Loads The MIC2169B supports pre-biased loads. Some applications have a pre-existing voltage on the output. This pre-existing or pre-biased load is generated by an external supply (other than the MIC2169B). During startup without pre-bias support, MIC2169A will pull the output voltage to ground through the inductor and low side FET (see figure 11). The MIC2169B prevents the current sinking of any preexisting voltage source at the output (see figure 12). It does this by keeping the low side FET off during the soft start period. In some applications this pre-bias current sink is not a problem, and the MIC2169A may be used. In some applications the pre-bias current sink may cause a problem, and the MIC2169B should be used. The MIC2169B can support up to 90% of a pre-bias condition (up to 90% of the final regulated output voltage) see figure 13.
Figure 12. MIC2169B startup with pre-bias support
Figure 12 shows MIC2169B startup with a pre-bias of 1V on the output, in which the pre-existing output voltage has no discharge.
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MIC2169B
Fig 13 MIC2169B startup with pre-bias support, pre-bias at 90% of Vout_final
Figure 13 shows MIC2169B startup with a pre-bias of 2.2V on the output (90% of VOUT) without the pre-existing output voltage discharge.
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MIC2169B
Inductor
Design and PCB Layout Guideline
Warning!!! To minimize EMI and output noise, follow these layout recommendations.
PCB Layout is critical to achieve reliable, stable and efficient performance. A ground plane is required to control EMI and minimize the inductance in power, signal and return paths. The following guidelines should be followed to insure proper operation of the MIC2169B converter.
IC
* Keep the inductor connection to the switch node (SW) short. * Do not route any digital lines underneath or close to the inductor. * Keep the switch node (SW) away from the feedback (FB) pin. * To minimize noise, place a ground plane underneath the inductor.
Output Capacitor
* Place the IC and MOSFETs close to the point of load (POL). * Use fat traces to route the input and output power lines. * Signal and power grounds should be kept separate and connected at only one location.
Input Capacitor
* Place the VIN input capacitor next. * Place the VIN input capacitors on the same side of the board and as close to the MOSFETs as possible. * Keep both the VIN and power GND connections short. * Place several vias to the ground plane close to the VIN input capacitor ground terminal. * Use either X7R or X5R dielectric input capacitors. Do not use Y5V or Z5U type capacitors. * Do not replace the ceramic input capacitor with any other type of capacitor. Any type of capacitor can be placed in parallel with the input capacitor. * If a Tantalum input capacitor is placed in parallel with the input capacitor, it must be recommended for switching regulator applications and the operating voltage must be derated by 50%. * In "Hot-Plug" applications, a Tantalum or Electrolytic bypass capacitor must be used to limit the overvoltage spike seen on the input supply with power is suddenly applied. * An additional Tantalum or Electrolytic bypass input capacitor of 22F or higher is required at the input power connection. * Use a 5 resistor from the input supply to the VDD pin on the MIC2169B. Also, place a 1F ceramic capacitor from this pin to GND, preferably not through a via. The capacitor must be located right at the IC. The VDD terminal is very noise sensitive and placement of the capacitor is very critical. Connections must be made with wide trace.
* Use a wide trace to connect the output capacitor ground terminal to the input capacitor ground terminal. * Phase margin will change as the output capacitor value and ESR changes. Contact the factory if the output capacitor is different from what is shown in the BOM. * The feedback trace should be separate from the power trace and connected as close as possible to the output capacitor. Sensing a long high current load trace can degrade the DC load regulation.
MOSFETs
* Low gate charge MOSFETs should be used to maximize efficiency, such as Si4800, Si4804BDY, IRF7821, IRF8910, FDS6680A and FDS6912A, etc.
RC Snubber
* Add a RC snubber of 1.4 resistor and a 1000pF capacitor from the switch node to ground pin. Place the snubber on the same side of the board and as close to the MOSFETs as possible. See page 8, Current Limiting section for more detail.
Schottky Diode (Optional)
* Place the Schottky diode on the same side of the board as the MOSFETs and VIN input capacitor. * The connection from the Schottky diode's Anode to the input capacitors ground terminal must be as short as possible. * The diode's Cathode connection to the switch node (SW) must be keep as short as possible.
Others
* Connect the current limiting (R2) resistor directly to the drain of top MOSFET Q1. * The feedback resistors R3 and R4/R5/R6 should be placed close to the FB pin. The top side of R3 should connect directly to the output node. Run this trace away from the switch node (junction of Q1, Q2, and L1). The bottom side of R3 should connect to the GND pin on the MIC2169B.
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Micrel, Inc. * The compensation resistor and capacitors should be placed right next to the COMP pin and the other side should connect directly to the GND pin on the MIC2169B rather than going to the plane.
MIC2169B * Add place holders for gate resistors on the top and bottom MOSFET gate drives. If necessary, gate resistors of 10 or less should be used.
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MIC2169B
Evaluation Board Schematics
R12 1.4 C15 1000p F
MIC2169B Evaluation Board Schematic
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MIC2169B
Bill of Materials
Item U1 Part Number MIC2169B-YMME Manufacturer Micrel, Inc. Description Buck controller Qty. 1
Q1, Q2 Q3 D1 D2
IRF7821-TR SI4390DY 2N7002E SD103BWS 1N5819HW SL04 CMMSH1-40 CDRH127LDNP-1R0NC HC5-1R0 SER1360-1R0 C3225X7R1C106M TPSD686M020R0070 594D686X0020D2T C2012X5R0J106M VJ1206Y104KXXAT TPSD337M006R0045
IR Vishay On Semiconductor Vishay Diodes Inc. Vishay Central Semi Sumida Cooper Electronic Coilcraft TDK AVX Vishay/Sprague TDK Vishay Victramon AVX Vishay/Sprague Vishay Dale
30V, N channel HEXFET , Power MOSFET 60V, N channel MOSFET 30V, Schottky Diode 40V, Schottky Diode
2 0 0 1 1 0 0 1 0 0 1 2 0 1 3 2 0 0 1 0 0 1 1 1 1 1 1 1 1 1 1 1 0 4
L1
1.0uH, 10A inductor
C1 C2, C3 C4 C5, C10, C12 C6, C7 C8 C9, C11 C13
10uF/16V, X7R Ceramic cap. 68uF, 20V Tantalum 10uF/6.3V, 0805 Ceramic cap. 0.1uF/25V Ceramic cap. 330uF, 6.3V, Tantalum Open open 1uF/16V, 0805 Ceramic cap.
C2012X7R1C105K GRM21BR71C105KA01B. VJ1206S105KXJAT VJ0603A102KXXAT VJ0603Y104KXXAT CRCW06034700JRT1 CRCW08051002FRT1 CRCW08053161FRT1 CRCW08054641FRT1 CRCW08051132FRT1 CRCW08051003FRT1 CRCW06034021FRT1 CRCW120610R0FRT1 CRCW12062R00FRT1 CRCW12061R40FRT1 2551-2-00-01-00-00-07-0
TDK muRata Vishay Victramon Vishay Victramon Vishay Victramon Vishay Vishay Vishay Vishay Vishay Vishay Vishay Vishay Vishay Vishay MilMax
C15 C16 C17 R2 R3 R4 R5 R6 R7 R8 R9 R10 R12 R14 J1, J3, J4, J5
Notes:
1000pF /25V, 0603 , NPO 0.1uF/25V Ceramic cap. open 470, 0603, 1/16W, 5%. 10k, 0805, 1/10W, 1% 3.16k, 0805, 1/10W , 1% 4.64k, 0805, 1/10W , 1% 11.3k, 0805, 1/10W, 1% 100k, 0805, 1/10W, 1% 4.02k, 0603 ,1/16W, 1% 10, 1/8W, 1206, 1% 2, 1/8 W, 1206, 1% 1.4, 1/8 W, 1206, 1% Open Turret Pins
1. 2. 3.
Micrel.Inc Vishay corp Diodes. Inc
408-944-0800 206-452-5664 805-446-4800
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4. 5. 6. 7. 8. 9. Sumida TDK muRata AVX International Rectifier Fairchild Semiconductor 408-321-9660 847-803-6100 800-831-9172 843-448-9411 847-803-6100 207-775-8100 561-752-5000 1-800-322-2645 631-435-1110
MIC2169B
10. Cooper Electronic 11. Coilcraft 12. Central Semi
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Micrel, Inc.
MIC2169B
MIC2169B PCB Layout
MIC2169B Top Layer
MIC2169B Bottom Layer
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MIC2169B
Package Information
10-Pin ePad MSOP (MME)
MICREL, INC. 2180 FORTUNE DRIVE SAN JOSE, CA 95131 USA
TEL +1 (408) 944-0800 FAX +1 (408) 474-1000 WEB http://www.micrel.com
The information furnished by Micrel in this data sheet is believed to be accurate and reliable. However, no responsibility is assumed by Micrel for its use. Micrel reserves the right to change circuitry and specifications at any time without notification to the customer. Micrel Products are not designed or authorized for use as components in life support appliances, devices or systems where malfunction of a product can reasonably be expected to result in personal injury. Life support devices or systems are devices or systems that (a) are intended for surgical implant into the body or (b) support or sustain life, and whose failure to perform can be reasonably expected to result in a significant injury to the user. A Purchaser's use or sale of Micrel Products for use in life support appliances, devices or systems is a Purchaser's own risk and Purchaser agrees to fully indemnify Micrel for any damages resulting from such use or sale. (c) 2009 Micrel, Incorporated.
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